Spectral Stitching Method to Increase Instantaneous Bandwidth in Vector signal Generators

ABSTRACT

Embodiments are described of devices and methods for processing a signal using a plurality of vector signal generators (VSGs). A digital signal may be provided to a plurality of signal paths, each of which may process a respective frequency band of the signal, the respective frequency bands having regions of overlap. The gain and phase of each signal path may be adjusted such that continuity of phase and magnitude are preserved through the regions of overlap. The adjustment of gain and phase may be accomplished by a complex multiply with a complex calibration constant. The calibration constant may be determined for each signal path by comparing the gain and phase of one or more calibration tones generated within each region of overlap. Each signal path may comprise a VSG to convert the respective signal to an analog signal, which may be combined to obtain a composite signal.

PRIORITY INFORMATION

This application is a continuation of U.S. patent application Ser. No.14/611,700, entitled “Spectral Stitching Method to IncreaseInstantaneous Bandwidth in Vector Signal Generators”, filed Feb. 2,2015, with inventors Stephen L. Dark et al., which is acontinuation-in-part application of U.S. patent application Ser. No.14/515,144, entitled “Spectral Stitching Method to IncreaseInstantaneous Bandwidth in Vector Signal Analyzers”, filed Oct. 15,2014, now U.S. Pat. No. 9,326,174 issued Apr. 26, 2016, with inventorsStephen L. Dark et al., each of which is hereby incorporated byreference as though fully and completely set forth herein.

FIELD OF THE INVENTION

The present invention relates to the field of signal processing, andmore particularly to systems and methods for increasing instantaneousbandwidth in a vector signal analyzer or a vector signal generator.

DESCRIPTION OF THE RELATED ART

Instantaneous bandwidth is an important banner specification for allradio frequency (RF) vector signal analyzers (VSAs) and RF vector signalgenerators (VSGs). The desire of the industry is to increase thebandwidth as much as possible without sacrificing dynamic range. In manycases, the limiting factor in achieving the largest possible bandwidthis the sample rate of the analog-to-digital converters (ADCs) anddigital-to-analog converters (DACs). While ADC and DAC vendors arealways working to increase the converter rates, there still exists adesire in many applications for bandwidths that exceed the capabilitiesof state-of-the-art ADCs and DACs. There are several different methodswithin the industry for achieving larger bandwidths, each with their owndisadvantages.

The prior methods for creating larger instantaneous bandwidths that donot intentionally sacrifice resolution fall into two categories: (1)Time-Interleaving (time) and (2) Quadrature Mixing (phase). Both ofthese methods are industry standards for increasing bandwidth.

Time-Interleaving uses converters that have larger bandwidths than theirsample rates allow. For ADCs, this is made possible by placing fastersample-and-hold circuits on the front-end than the ADC backend iscapable of digitizing. Then, by taking N ADCs and staggering them intime by the ADC sample period divided by N, the samples can beinterleaved together to create an effective larger sample rate. Thismethod introduces errors resulting from inaccuracies in staggering thetime alignment and from differences in the magnitude and phase betweenthe various ADCs. Therefore, several online and offline DSP correctionalgorithms have been created to combat these effects. In general, it isdifficult to achieve more than 8 bits of dynamic range without DSPcorrection. With offline DSP correction, this can be improved to betterthan 12 bits of dynamic range but can be very sensitive to temperature.Online methods typically have to assume something about the inputsignal, and other negative effects occur when those assumptions arebroken.

Quadrature mixing uses a quadrature down-converter or up-converter tomix an RF signal into or from two signals, an in-phase signal and aquadrature-phase signal. In the case of a down-converter, the RF signalis mixed with a sinusoid to create the in-phase signal and the same RFsignal is mixed with a sinusoid that is 90 degrees out of phase with thein-phase sinusoid to create the quadrature phase signal. Finally, eachof these two analog signals is digitized with ADCs. These two signalsare typically represented as a single complex signal, where the in-phasesignal represents the real part and the quadrature-phase part of thesignal represents the imaginary part. As a result, the positivefrequency bandwidth is independent of the negative frequency bandwidth.Thus the net effect is a doubling of the bandwidth. Using this method,the full resolution of the data converters is preserved. However, thismethod typically creates a DC leakage spur and an image spur. Inaddition, the method only scales to two converts.

Thus, there exists a need for mechanisms capable of achieving the goalof larger instantaneous or modulation bandwidths from smaller bandwidthswithout the scalability and image rejection issues of the quadraturemixing technique and without the inaccuracies present for timeinterleaving methods.

SUMMARY

Methods and systems are disclosed for processing a digital signal usinga plurality of parallel processing paths. In a presented method, each ofthe plurality of parallel processing paths may be provided with arespective component signal comprising a copy of a respective frequencyband of an input signal. The combination of the respective frequencybands may comprise an aggregate frequency band having an aggregatecenter frequency. Each respective frequency band may have a respectiveregion of overlap with at least one other respective frequency band.Each respective frequency band may also have a respective centerfrequency with a respective frequency offset from the aggregate centerfrequency. At least a portion of each of the parallel processing pathsmay be phase-locked and time-synchronized with respect to the otherparallel processing paths.

Each of the parallel processing paths may process the providedrespective component signal. The processing may comprisefrequency-shifting, filtering, and adjusting the gain and phase of therespective component signal, and converting the respective componentsignal to an analog signal. The frequency-shifting may comprise shiftingthe respective component signal such that the respective centerfrequency is shifted to baseband. The filtering the respective componentsignals may be configured to cause a sum of the component signals tohave a unity frequency response within each region of overlap. Thefiltering may be performed using a digital half-band filter. Theadjusting gain and phase of the respective component signals may beconfigured to cause the sum of the component signals to have acontinuous frequency response over the aggregate frequency band. Theconverting the respective component signal to an analog signal maycomprise using a respective vector signal generator associated with therespective parallel processing path. The converting may comprise furtherfrequency-shifting the respective component signals to cause theaggregate center frequency to be located at a desired carrier frequency,and each respective center frequency to be offset from the desiredcarrier frequency by the respective frequency offset.

Optionally, the processing may include additional steps, such asdecimating the respective component signal to a rate that is less thanor equal to a maximum sample rate of the respective vector signalgenerator.

The method may further comprise combining the respective componentsignals to obtain a composite signal.

An apparatus is presented for processing a digital signal. The apparatusmay comprise a digital signal processor comprising a plurality ofparallel processing paths. The digital signal processor may beconfigured to provide to each of the parallel processing paths arespective component signal comprising a copy of at least a respectivefrequency band of the digital signal. Each respective frequency band mayhave a respective center frequency and a respective region of overlapwith at least one other respective frequency band. The combination ofthe respective frequency bands may comprise an aggregate frequency bandhaving an aggregate center frequency, wherein the respective centerfrequency of each respective frequency band has a respective frequencyoffset from the aggregate center frequency. At least a portion of eachof the parallel processing paths may be phase-locked andtime-synchronized with respect to the other parallel processing paths.Each of the parallel processing paths may be configured tofrequency-shift, filter, and adjust the gain and phase of the respectivecomponent signal, as discussed above. Optionally, each of the parallelprocessing paths may be further configured to perform additional steps,such as decimating the respective component signal to a rate that isless than or equal to a maximum sample rate of the respective vectorsignal generator.

The apparatus may further comprise a plurality of output ports, eachconfigured to provide to a respective vector signal generator (VSG) anoutput of one of the plurality of parallel processing paths. Theapparatus may further comprise a plurality of input ports, eachconfigured to receive from a respective VSG a respective analog signalcomprising an analog version of the output of the respective one of theplurality of output ports. Each respective analog signal received at oneof the input ports may have been frequency-shifted relative to therespective output of the respective output port, such that the aggregatecenter frequency is located at a desired carrier frequency, and eachrespective center frequency is offset from the desired carrier frequencyby the respective frequency offset. The apparatus may further comprise asignal combiner configured to combine the respective analog signals toobtain a composite signal.

A method is presented for calibrating a signal processing systemincluding at least a first vector signal generator (VSG) and a secondVSG. The method may comprise providing to a first signal processing pathcomprising the first VSG a first digital component signal comprising afirst frequency band within an aggregate frequency band of a digitalinput signal. The aggregate frequency band may have an aggregate centerfrequency, and the first frequency band may have a first centerfrequency at a first frequency offset from the aggregate centerfrequency. The method may further comprise providing to a second signalprocessing path comprising the second VSG a second digital componentsignal comprising a second frequency band within the aggregate frequencyband of the digital input signal. The second frequency band may have asecond center frequency at a second frequency offset from the aggregatecenter frequency. The second frequency band may also have a region ofoverlap with the first frequency band, the region of overlap containinga calibration tone. The second VSG may be phase-locked andtime-synchronized with respect to the first VSG.

The method may further comprise frequency shifting each of the first andsecond component signals, such that the first center frequency and thesecond center frequency are each shifted to baseband. The method mayfurther comprise converting the first and second digital componentsignals to first and second analog component signals, respectively,using the first VSG and the second VSG. Converting the first and seconddigital component signals may comprise shifting each of the first andsecond component signals, such that the aggregate center frequency islocated at a desired carrier frequency, and each respective centerfrequency is offset from the desired carrier frequency by the respectivefrequency offset.

The method may further comprise computing a complex calibration constantbased on a magnitude ratio and a phase difference. The magnitude ratiomay be determined by a magnitude of the calibration tone as output bythe first VSG and a magnitude of the calibration tone as output by thesecond VSG. The phase difference may be determined by a phase of thecalibration tone as output by the first VSG and a phase of thecalibration tone as output by the second VSG. For example, computing thecomplex calibration constant may comprise disabling the output of thesecond VSG, measuring the magnitude and phase of the calibration tone asoutput by the first VSG while the output of the second VSG is disabled,disabling the output of the first VSG and enabling the output of thesecond VSG after said measuring, and measuring the magnitude and phaseof the calibration tone as output by the second VSG while the output ofthe first VSG is disabled. As another example, computing the complexcalibration constant may comprise iteratively adjusting the magnitudeand phase of the second VSG to cause deconstructive interference betweenthe calibration tone as output by the first VSG and the calibration toneas output by the second VSG, such that the total power at the frequencyof the calibration tone as output by the first VSG and the second VSG isminimized; and computing the complex calibration constant as theadjustment made to the magnitude and phase of the second VSG, rotated by180 degrees. As another example, two calibration tones may be atdifferent frequencies but with a known relationship. In this case boththe first VSG and the second VSG may each generate a respectivecalibration tone at the same time in a synchronized manner, and themagnitude and phase differences may be simultaneously sampled. Thecomplex calibration constant may be computed by taking intoconsideration the known relationship between the calibration tones.

The method may further comprise storing the complex calibration constantin memory, wherein the complex calibration constant is useable tocorrect phase and gain mismatch between the first VSG and the secondVSG. The first VSG may comprise a first local oscillator and the secondVSG comprises a second local oscillator. Optionally, the phases of thefirst local oscillator and the second local oscillator may bedeterministic, such that the stored complex calibration constant isuseable to correct phase and gain mismatch between the first VSG and thesecond VSG without recalibration, even after phase lock between thefirst VSG and the second VSG is lost and reacquired.

Optionally, the method may include additional steps, such as filteringeach of the first and second component signals, and/or decimating eachof the first and second component signals, as discussed above.

BRIEF DESCRIPTION OF THE DRAWINGS

A better understanding of the present inventions can be obtained whenthe following detailed description is considered in conjunction with thefollowing drawings:

FIG. 1 is a block diagram illustrating an embodiment of a system forperforming spectral stitching in a receive path;

FIG. 2 is a block diagram illustrating an embodiment of parallel vectorsignal analyzers (VSAs);

FIGS. 3a, 3b, and 3c illustrate exemplary embodiments of respectivefrequency bands within an aggregate frequency band with and withoutcalibration tones;

FIG. 4 illustrates the signal response of a half-band filter;

FIG. 5 is a block diagram illustrating another embodiment of a systemfor performing spectral stitching in a receive path;

FIGS. 6a and 6b illustrate measurements of a calibration tone, asperformed by two VSAs before and after phase and magnitude adjustment,represented in the time domain;

FIG. 7 is a block diagram illustrating an embodiment of a system forperforming spectral stitching in a transmit path; and

FIG. 8 is a block diagram illustrating an embodiment of a digital signalprocessing block for use in the system of FIG. 7.

While the invention is susceptible to various modifications andalternative forms, specific embodiments thereof are shown by way ofexample in the drawings and are herein described in detail. It should beunderstood, however, that the drawings and detailed description theretoare not intended to limit the invention to the particular formdisclosed, but on the contrary, the intention is to cover allmodifications, equivalents and alternatives falling within the spiritand scope of the present invention as defined by the appended claims.Note that the various section headings in the following DetailedDescription are for organizational purposes only and are not meant to beused to limit the claims.

DETAILED DESCRIPTION

Incorporation by Reference

The following references are incorporated by reference as if fully andcompletely disclosed herein:

U.S. Patent Application No. 2013/0343490, filed Jun. 20, 2012, entitled“Synchronizing Receivers in a Signal Acquisition System”, invented byWertz et al.;

U.S. Pat. No. 7,624,294, issued on Nov. 24, 2009, entitled“Synchronizing Measurement Devices Using Trigger Signals”, invented byCraig M. Conway; and

U.S. Pat. No. 7,315,791, issued on Jan. 1, 2008, entitled “ApplicationProgramming Interface for Synchronizing Multiple InstrumentationDevices”, invented by Kosta Ilic et al.

Terminology

The following is a glossary of terms used in the present application:

Memory Medium—Any of various types of memory devices or storage devices.The term “memory medium” is intended to include an installation medium,e.g., a CD-ROM, floppy disks 105, or tape device; a computer systemmemory or random access memory such as DRAM, DDR RAM, SRAM, EDO RAM,Rambus RAM, etc.; a non-volatile memory such as a Flash, magnetic media,e.g., a hard drive, or optical storage; registers, or other similartypes of memory elements, etc. The memory medium may comprise othertypes of memory as well or combinations thereof. In addition, the memorymedium may be located in a first computer in which the programs areexecuted, or may be located in a second different computer whichconnects to the first computer over a network, such as the Internet. Inthe latter instance, the second computer may provide programinstructions to the first computer for execution. The term “memorymedium” may include two or more memory mediums which may reside indifferent locations, e.g., in different computers that are connectedover a network.

Programmable Hardware Element—includes various hardware devicescomprising multiple programmable function blocks connected via aprogrammable interconnect. Examples include FPGAs (Field ProgrammableGate Arrays), PLDs (Programmable Logic Devices), FPOAs (FieldProgrammable Object Arrays), and CPLDs (Complex PLDs). The programmablefunction blocks may range from fine grained (combinatorial logic or lookup tables) to coarse grained (arithmetic logic units or processorcores). A programmable hardware element may also be referred to as“reconfigurable logic”.

Computer System—any of various types of computing or processing systems,including a personal computer system (PC), mainframe computer system,workstation, network appliance, Internet appliance, personal digitalassistant (PDA), television system, grid computing system, or otherdevice or combinations of devices. In general, the term “computersystem” can be broadly defined to encompass any device (or combinationof devices) having at least one processor that executes instructionsfrom a memory medium.

Local Oscillator (LO)—a circuit configured to generate a periodic signalat a specified frequency and amplitude. The periodic signal may be apure sinusoid, and its frequency and/or amplitude may be programmable.The periodic signal may or may not be phase or frequency locked toanother periodic signal.

Overview

Time interleaving uses time as the mechanism to increase the bandwidthwhile quadrature mixing uses phase as its mechanism. The present“spectral stitching” approach uses frequency as its mechanism to achievelarger instantaneous bandwidths. The spectral stitching approach may beapplied to both signal receivers, such as RF (radio frequency)receivers, for example, and signal generators, such as RF generators,for example, as discussed below. As used herein, the term “RF” isintended to include the full spectrum of communication frequencies, andincludes at least radio and microwave frequencies.

Embodiments of the present invention may be realized in any of variousforms. For example, in some embodiments, the present invention may berealized as a computer-implemented method, a computer-readable memorymedium, or a computer system. In other embodiments, the presentinvention may be realized using one or more custom-designed hardwaredevices such as ASICs. In other embodiments, the present invention maybe realized using one or more programmable hardware elements such asFPGAs.

In some embodiments, a computer-readable memory medium may be configuredso that it stores program instructions and/or data, where the programinstructions, if executed by a computer system, cause the computersystem to perform a method, e.g., any of a method embodiments describedherein, or, any combination of the method embodiments described herein,or, any subset of any of the method embodiments described herein, or,any combination of such subsets.

In some embodiments, a computer system may be configured to include aprocessor (or a set of processors) and a memory medium, where the memorymedium stores program instructions, where the processor is configured toread and execute the program instructions from the memory medium, wherethe program instructions are executable to implement any of the variousmethod embodiments described herein (or, any combination of the methodembodiments described herein, or, any subset of any of the methodembodiments described herein, or, any combination of such subsets). Thecomputer system may be realized in any of various forms. For example,the computer system may be a personal computer (in any of its variousrealizations), a workstation, a computer on a card, anapplication-specific computer in a box, a server computer, a clientcomputer, a hand-held device, a tablet computer, a wearable computer,etc.

Receive Path

In a receive path, spectral stitching may be performed by using aplurality N of vector signal analyzers (VSAs) to digitize an analoginput receive (RX) signal, such as an RF signal, where each VSA handlesa respective frequency band of the signal. Together the respectivefrequency bands comprise an aggregate frequency band of interest. Theoutputs of the N VSAs may therefore be recombined to form a compositesignal having a bandwidth on the order of N times the bandwidth of eachindividual VSA, thus covering the entire aggregate frequency band. Theaggregate frequency band may be a region of interest within the input RXsignal.

FIG. 1 illustrates a block diagram of an embodiment of a system 100 forperforming spectral stitching in a signal path receiving an inputRXsignal. As shown in FIG. 1, the system 100 includes three VSAs 108 a-c.Other embodiments may include another number N of VSAs. It should beappreciated that the terms “VSA” and “vector signal analyzer,” as usedherein, may encompass any type of signal analyzer, digitizer, receiver,or other device capable of converting, or configured to convert, ananalog input signal to a digital output signal.

One or more calibration tones may be added to the inputRX signal to aidin calibrating the plurality of VSAs, as discussed below. In someembodiments, the one or more calibration tones may be generated bycalibration tone generator 104, and added to the input signal by a powercombiner 102. In other embodiments, the one or more calibration tonesmay be added to the input signal using a simple two-port mux, or usingany other signal combining technique known in the art. In someembodiments, the system 100 may include multiple calibration tonegenerators, which may require the power combiner 102 to have more thantwo inputs. The one or more calibration tones may be generated and addedto the input signal while the system 100 is operating in a calibrationmode, as discussed below. Thus, in normal operation, the power combiner102 may not add the one or more calibration tones to the input signal.

A copy of at least a portion of the inputRX signal may be provided toeach of the VSAs 108 a-c. This may be accomplished by splitting theinput signal, such as by using a power splitter 106, having N outputs.

The output of each VSA 108 a-c may be provided to a summing unit 110.The output of the summing unit 110 is a composite signal representingthe sum of the output of each of the VSAs 108 a-c.

As mentioned previously, each of the VSAs 108 a-c may handle a differentfrequency band of the inputRX signal. To ensure continuity across thefull aggregate frequency band, the different frequency bands handled bythe respective VSAs 108 a-c should overlap. Thus, in order for the sumof the outputs of the VSAs 108 a-c to accurately represent a digitizedversion of the aggregate frequency band of the inputRX signal, theoutputs of the VSAs 108 a-c may be further processed to providecontinuity through the regions of overlap. Each of the VSAs 108 a-c maytherefore comprise signal processing capabilities beyond thosetraditionally included in a VSA. In some embodiments, each of the VSAs108 a-c may not comprise a stand-alone VSA. For example, each of theVSAs 108 a-c may be implemented as a signal processing path on aprogrammable hardware element, multiprocessor system, etc.

FIG. 2 illustrates a block diagram providing further detail of anembodiment of the VSAs 108 a-c. Specifically, blocks 202 a-210 arepresent details of an embodiment of VSA 108 a, blocks 202 b-210 brepresent details of an embodiment of VSA 108 b, and blocks 202 c-210 crepresent details of an embodiment of VSA 108 c. Summing unit 110 isalso included in FIG. 2 for context.

At each of the digitize blocks 202 a-c, a component signal comprising arespective frequency band of the inputRX signal may be digitized. Eachof the digitize blocks 202 a-c may include functionality included in atraditional VSA.

In some embodiments, each of the digitize blocks 202 a-c may receive acopy of the entire inputRX signal, e.g. from the signal splitter 106. Inother embodiments, each of the digitize blocks 202 a-c may receive onlya respective portion of the inputRX signal. Because each of the VSAs 108a-c may handle a different frequency band of the inputRX signal, each ofthe digitize blocks 202 a-c may digitize a respective component signalcomprising a respective frequency band, each respective frequency bandidentified by a respective center frequency.

FIG. 3a illustrates an exemplary embodiment of frequency bands asreceived by the digitize blocks 202 a-c, represented in the frequencydomain. In this example, frequency band 302 represents the frequencyband of the digitize block 202 a (i.e. VSA 108 a), frequency band 304represents the frequency band of the digitize block 202 b (i.e. VSA 108b), and frequency band 306 represents the frequency band of the digitizeblock 202 c (i.e. VSA 108 c). The region covered by frequency bands304-306 together represents the aggregate frequency band identified byan aggregate center frequency. Each respective center frequency isoffset from the aggregate center frequency by a respective frequencyoffset. In some circumstances a respective center frequency may beoffset from the aggregate center frequency by 0 Hz, as in the example offrequency band 304.

The frequency bands may overlap to avoid gaps within the aggregatefrequency band. For example, the region 308 represents a region ofoverlap between frequency band 302 and frequency band 304 (i.e. betweenthe respective component signals of VSAs 108 a and 108 b), and theregion 310 represents the region of overlap between frequency band 304and frequency band 306 (i.e. between the respective component signals ofVSAs 108 b and 108 c).

The digitizing performed by each of the digitize blocks 202 a-c maycomprise performing I/Q demodulation on the respective component signalto produce a pair of analog I (in-phase) and Q (quadrature) signals.

The digitizing performed by each of the digitize blocks 202 a-c may alsocomprise frequency-shifting the respective component signal (or the I/Qsignal pair) such that the respective center frequency is shifted tobaseband. Each of the digitize blocks 202 a-c may then filter outportions of the shifted signal that are outside the respective frequencyband, e.g. by using a low-pass filter. Alternatively, each of thedigitize blocks 202 a-c may frequency-shift the received inputRX signalto a position other than baseband (or forego frequency-shifting), andfilter the shifted signal using a band-pass filter.

In one embodiment, each of the digitize blocks 202 a-c may comprise arespective local oscillator (LO) operating at the respective centerfrequency. The respective LO may be used, for example, infrequency-shifting the respective center frequency to baseband. The LOsof the digitize blocks 202 a-c may have a fixed phase differencerelative to each other. For example, the LOs may be locked to a commonreference, such that the relative phases between the devices will remainfixed.

The digitizing performed by each of the digitize blocks 202 a-c mayfurther comprise complex sampling the filtered signal, as known in theart. The VSAs 108 a-c may be time-synchronized, such that the respectivesignals may be sampled simultaneously in each of the digitize blocks 202a-c. Alternatively, the respective signals may be sampled at aconsistent offset of time. In this case, the consistent offset may bemeasured and corrected. Each of the digitize blocks 202 a-c may output acomplex (I/Q signal.

At filter blocks 204 a-c, the respective component signals may befiltered. Because the respective frequency bands of the respectivecomponent signals overlap in frequency, as shown in FIG. 3, the overlapregions should be filtered to prevent power spikes, or other artificialincreases in magnitude, in the overlap regions when the respectivesignals are summed by summing unit 110. In other words, the respectivecomponent signals should be filtered such that their sum appearscontinuous. Specifically, the respective component signals may befiltered such that the sum of overlapping signals provides a unityresponse at all points within the aggregate frequency band. Moregenerally, this continuous-sum filtering may be configured in any mannersuch that the summed signals approximate the result that would beachieved if the entire aggregate frequency band had been digitized by asingle VSA having sufficient bandwidth to digitize the entire aggregatefrequency band.

Various filter shapes may be used to accomplish this. For example, FIG.4 illustrates the response of a half-band filter, where the solid trace402 represents a filter response for a first VSA and the dotted trace404 represents a filter response for a second VSA with an overlappingfrequency band. In FIG. 4, the crossover point is located at 30 MHzwhere there is a 10 MHz crossover region. While a half-band filterinherently has the needed spectral characteristics to filter overlappingfrequency bands to sum together to produce unity gain, it forces thecrossover point to occur at the sampling frequency divided by four,fs/4. In other embodiments, this crossover may be moved further out infrequency using other filter methods, e.g., to increase theeffectiveness of the natural instantaneous bandwidth of each device.

The filtering illustrated as filter blocks 204 a-c may happen at any ofvarious points in the VSA. For example, the filtering may occur afterthe interpolate blocks 206 a-c. Alternatively, some embodiments mayperform the filtering of the filter blocks 204 a-c within the digitizeblocks 202 a-c, e.g., concurrently with the low-pass filtering of thedigitize blocks 202 a-c. In this case, the filtering may be performed byan analog filter prior to complex sampling of the filtered signal.

At interpolate blocks 206 a-c, each respective component signal may beinterpolated. The interpolation factor should be set such that eachrespective interpolate block 206 interpolates the respective componentsignal to at least the effective I/Q rate required for the “stitched”data's bandwidth. For example, in one embodiment, the effective I/Q ratemay be required to be at least the Nyquist rate of the respectivecomponent signal. In another embodiment, a higher rate (e.g., 1.25 timesthe Nyquist rate) may be selected.

At frequency shift blocks 208 a-c, each respective component signal maybe shifted into the proper location in frequency relative to the otherdevices. As a result, each device will frequency shift its interpolatedspectrum to a different location. Specifically, each respectivecomponent signal may be shifted such that its respective centerfrequency is offset from baseband by the respective frequency offset bywhich it was originally offset from the aggregate center frequency.Thus, the entire aggregate frequency band is frequency shifted to centerat baseband.

For example, in the case that there are three VSAs each using half-bandfilters with an I/Q rate of 120 MHz, then the cross-over points will belocated at positive and negative 30 MHz. This means that the threerespective center frequencies may be shifted to [−60 MHz, 0 Hz, 60 MHz].Thus, the respective frequency bands should be defined such that therespective frequency offsets are [−60 MHz, 0 Hz, 60 MHz] relative to theaggregate center frequency.

If the digitize blocks 202 a-c previously shifted the respective centerfrequencies to baseband, then VSA 108 a may, in this example, frequencyshift its spectrum to the left by 60 MHz, VSA 108 b may shift by 0 Hz,and VSA 108 c may frequency shift its spectrum to the right by 60 MHz.In other words, each respective center frequency may be shifted by itsrespective frequency offset. In embodiments in which the respectivefrequencies were shifted by the digitize blocks 202 a-c to a frequencyother than baseband, then the respective center frequencies may beshifted by some value other than the respective frequency offsets.

At the gain and phase correction blocks 210 a-c, the magnitude and phaseof each respective component signal may be adjusted to make the spectrumcontinuous through the regions of overlap. This gain and phasecorrection may comprise a complex multiply of each of one or more of therespective component signals with a respective calibration constant.Determining a calibration constant for each of the gain and phasecorrection blocks 210 a-c is discussed below.

Where the VSAs 108 a-c are not time-synchronized, but have a constantdelay relative to each other, the gain and phase correction blocks 210a-c may be further configured to measure and correct the delay.

FIG. 5 illustrates a block diagram of an embodiment of a second system500 for performing spectral stitching in a signal path receiving aninputRX signal. As shown in FIG. 5, the system 500 includes three VSAs508 a-c. Other embodiments may include another number N of VSAs.

In FIG. 5, the power combiner 102, the calibration tone generator 104,and the power splitter 106 may operate as described with regard toFIG. 1. The VSAs 508 a-c may be standard VSAs as known in the art,without the additional signal processing capabilities of VSAs 108 a-c.Instead, the additional signal processing functions may be performed bya separate digital signal processing block 510. For example, the digitalsignal processing block 510 may perform the functions of filtering,interpolating, frequency shifting, and gain and phase correction, asdiscussed with regard to FIG. 2, blocks 204-210, for each of the VSAs508 a-c. For example the digital signal processing block 510 maycomprise a separate signal processing path for processing the output ofeach of the VSAs 508 a-c. The digital signal processing block 510 mayfurther comprise a summing function, which may function in a mannersimilar to the summing unit 110.

The system 500 presents an advantage over the system 100, in that thesystem 500 may allow a user to utilize standard, off-the-shelf VSAs. Forexample, system 500 may be realized in the form of a signal processingchassis comprising the digital signal processing block 510, andoptionally further comprising one or more of the power combiner 102, thecalibration tone generator 104, and the power splitter 106. The signalprocessing chassis may further comprise slots to accept a plurality ofVSAs, which may be standard, off-the-shelf VSAs. The signal processingchassis may be configured to operate with a variable number of VSAs,according to the preferences of the user. Further, the VSAs may differin bandwidth, resolution, or other characteristics, according to thepreferences of the user.

Determining VSA Calibration Constants

In order to adjust the magnitudes and phases of the respective signalsin the receive path to provide continuity through the regions ofoverlap, the relative magnitudes and phases between the respectivesignals without adjustment may be determined. This may be performed byinjecting a calibration tone at each crossover point, or region ofoverlap, of the respective frequency bands. The calibration tone maythen be measured and compared by the VSAs. Differences and/or ratiosbetween the measurements by different VSAs of the magnitude and phase ofthe calibration tone may be used to determine calibration constants foreach of the VSAs.

For example, to determine calibration constants for a system such as thesystem 100 shown in FIGS. 1-2, or the system 500 shown in FIG. 5, thesystem may be set to a calibration mode. The calibration mode will bediscussed herein with respect to the system 100 shown in FIGS. 1-2.However, the same principles may be applied to other embodiments, suchas the system 500.

As shown in FIG. 1, the power combiner 102 may be used to combine theinput signal with one or more calibration tones from the calibrationtone generator 104. A calibration tone may comprise a real tone at asingle known frequency falling within a region of overlap. A calibrationtone may also have a known magnitude. In some embodiments, a calibrationtone may be injected at a single region of overlap at a time, with eachregion of overlap being treated sequentially. In other embodiments,calibration tones may be injected at multiple, or all, regions ofoverlap simultaneously. The combined signal may then be used as theinput to the splitter 106 where each of the splitter's outputs may bethe input to one of N VSAs, such as VSAs 108 a-c.

At the N VSAs, each of the N-1 regions of overlap may comprise arespective calibration tone, as shown in FIG. 3b (either simultaneouslyor sequentially). As illustrated, calibration tone 312 may be generatedwithin region of overlap 308 (i.e. within the respective signals of VSAs108 a and 108 b), and calibration tone 314 may be generated withinregion of overlap 310 (i.e. within the respective signals of VSAs 108 band 108 c).

At blocks 202-208, each respective signal may be digitized, filtered,interpolated, and frequency shifted, as discussed above with regard toFIG. 2. However, the gain and phase correction block 210 may operatedifferently in the calibration mode. Specifically, the gain and phasecorrection block 210 may measure the respective calibration tone withineach region of overlap. For example, the gain and phase correction block210 a (of VSA 108 a) may measure the calibration tone 312, since itfalls within the respective frequency band 302, which is processed bythe VSA 108 a. The gain and phase correction block 210 b (of VSA 108 b)may also measure the calibration tone 312, since it also falls withinthe respective frequency band 304, which is processed by the VSA 108 b.

FIG. 6a illustrates exemplary results of measurements of the calibrationtone 312, as performed by the VSAs 108 a and 108 b, represented in thetime domain. As illustrated, the in-phase (I) and quadrature (Q)components of the calibration tone 312 as measured by the VSA 108 a havea phase and magnitude that are different from the phase and magnitude ofthe I and Q components of the calibration tone 312 as measured by theVSA 108 b. This may result from normal differences in the hardware,temperature, etc. of the VSAs 108 a and 108 b. In this condition, thesum of the respective signals output by the VSAs 108 a and 108 b willnot be continuous through the region of overlap, because of the mismatchin phase and magnitude.

Using the measurements of the calibration tone within each region ofoverlap, respective calibration constants may be determined, for use inrealigning the respective signals through each region of overlap. Forexample, for each respective VSA, a complex calibration constant may bedetermined that, when complex multiplied by a calibration tone measuredby that respective VSA, will result in an output calibration tone havinga phase and magnitude matching an output calibration tone of an adjacentVSA. FIG. 6b illustrates output calibration tones of VSAs 108 a and 108b. The output calibration tones shown in FIG. 6b represent the signalsshown in FIG. 6a after being multiplied by the determined calibrationconstants.

In one embodiment, the magnitudes of the calibration tones generated bythe calibration tone generator 104 may be known. The calibrationconstants may therefore be determined such that the output calibrationtones have a magnitude matching the generated calibration tones. Inanother embodiment, the calibration constants may merely be determinedsuch that the output calibration tones produced by adjacent VSAs havethe same magnitude.

For example, calibration tone 312 may be measured by both VSA 108 a andVSA 108 b. Calibration tone 314 may be measured by both VSA 108 b andVSA 108 c. The output signal of VSA 108 a will include an outputcalibration tone corresponding to calibration tone 312. The outputsignal of VSA 108 b will include output calibration tones correspondingto calibration tone 312 and calibration tone 314. The output signal ofVSA 108 c will include an output calibration tone corresponding tocalibration tone 314.

A first calibration constant may optionally be determined for VSA 108 asuch that complex multiplication of the first calibration constant withthe output signal of VSA 108 a results in the output calibration tonecorresponding to calibration tone 312 having a magnitude matching theknown magnitude of calibration tone 312.

A second calibration constant may be determined for VSA 108 b such thatcomplex multiplication of the second calibration constant with theoutput signal of VSA 108 b results in the output calibration tonecorresponding to calibration tone 312 having a phase matching the phaseof the output calibration tone of VSA 108 a. The second calibrationconstant may further be determined such that the output calibration tonecorresponding to calibration tone 312 has a magnitude matching the knownmagnitude of calibration tone 312 and/or the magnitude of the outputcalibration tone of VSA 108 a corresponding to calibration tone 312. Inone embodiment, a calibration constant determined to cause the outputcalibration tone of 108 b to match the phase and magnitude of the outputcalibration tone of VSA 108 a may be determined by performing a complexdivision of the calibration tone 312 as measured by VSA 108 a by thecalibration tone 312 as measured by VSA 108 b. Because the complexmultiplication of the second calibration constant is performed with theentire output signal of VSA 108 b, the output calibration tone of VSA108 b corresponding to calibration tone 314 is also adjusted.

A third calibration constant may be determined for VSA 108 c such thatcomplex multiplication of the third calibration constant with the outputsignal of VSA 108 c results in the output calibration tone correspondingto calibration tone 314 having a phase matching the phase of theadjusted output calibration tone of VSA 108 b corresponding tocalibration tone 314. The third calibration constant may further bedetermined such that the output calibration tone corresponding tocalibration tone 314 has a magnitude matching the known magnitude ofcalibration tone 314, and/or the magnitude of the output calibrationtone of VSA 108 b corresponding to calibration tone 314.

The calibration constants for each of the VSAs should be determined, orre-determined, each time the phases of the VSAs change relative to eachother. For example, the relative phases of the VSAs may change if therelative phases of the LOs of the VSAs change. This may occur, e.g., ifthe LO frequencies of one or more VSAs change and the one or more VSAsare relocked. In some VSAs, the phase of the LO can be made to bedeterministic. If this is the case, then a calibration constant may bedetermined once and stored for each frequency of the VSA. The storedcalibration constant may then be recalled at some future time withoutthe need for recalibration.

Transmit Path

In a transmit path, spectral stitching may be performed by using aplurality N of vector signal generators (VSGs) to generate an outputanalog transmit (TX) signal, such as an RF signal, where each VSGhandles a respective frequency band of the signal. Together therespective frequency bands comprise an aggregate frequency band ofinterest. The outputs of the N VSAs may therefore be combined to form acomposite signal having a bandwidth on the order of N times thebandwidth of each individual VSG, thus covering the entire aggregatefrequency band.

FIG. 7 illustrates a block diagram of an embodiment of a system 700 forperforming spectral stitching in a signal path generating an outputanalog signal. As shown in FIG. 7, the system 700 includes three VSGs704 a-c. Other embodiments may include another number N of VSGs. Itshould be appreciated that the terms “VSG” and “vector signalgenerator,” as used herein, may encompass any type of signal generator,transmitter, or other device capable of converting, or configured toconvert, a digital input signal to an analog output signal.

A digital input signal, illustrated in FIG. 7 as the “TX Signal”, may beprovided to the system. The input TX Signal may comprise a complexdigital signal having a bandwidth that is larger than the bandwidth ofeach respective VSG. Therefore, each of the VSGs 704 a-c may be providedwith a respective component signal comprising at least a portion of theinput TX Signal. Specifically, each component signal may comprise arespective frequency band of the input TX Signal, and each of the VSGs704 a-c may process the respective frequency band. Each respectivefrequency band may have a respective center frequency having arespective frequency offset from an aggregate center frequency of theaggregate frequency band.

To ensure continuity across the full aggregate frequency band, therespective frequency bands should overlap. Thus, in order for theoutputs of the VSGs 704 a-c to be recombined to accurately represent ananalog version of the aggregate frequency band of the digital inputsignal, the component signals may be further processed to providecontinuity through the regions of overlap. Each of the VSGs 704 a-c maytherefore comprise signal processing capabilities beyond thosetraditionally included in a VSG. Alternatively, such further processingmay be performed by signal processing circuitry outside of the VSGs 704a-c, such as by the digital signal processing (DSP) block 702, which isdescribed more fully below. Such an embodiment would allow a user toimplement the present invention using standard off-the-shelf VSGs. Inother embodiments, each of the VSGs 704 a-c may not comprise astand-alone VSG. For example, each of the VSGs 704 a-c may beimplemented as a signal processing path on a programmable hardwareelement, multiprocessor system, etc.

Once the further processing has been performed, e.g. by the DSP block702 or by each of the VSGs 704 a-c., each of the VSGs 704 a-c mayconvert the respective component signal to an analog signal. Each of theVSGs 704 a-c may also up-convert the respective component signal suchthat the aggregate center frequency is located at a desired carrierfrequency, and each respective center frequency is offset from thedesired carrier frequency by the respective frequency offset.

The combiner 706 may comprise a power combiner with a plurality ofinputs, or may comprise any other hardware for combining analog signals,as known in the art. The combiner 706 may receive as inputs the outputsof the VSGs 704 a-c, and may output a composite signal comprising acombination of its inputs.

A power splitter 708 may provide copies of the composite signal as anoutput of the system, i.e. as “TX Out” shown in FIG. 7, and also to acalibration receiver 710. The calibration receiver 710 may be used toreceive one or more calibration tones for use in determining calibrationconstants for one or more of the VSGs 704 a-c, as discussed below. Thecalibration receiver 710 may be phase locked to the VSGs 704 a-c.

FIG. 8 is a block diagram illustrating the DSP block 702 in greaterdetail. The DSP block 702 may comprise a plurality of parallelprocessing paths, each of which may process one of the respectivecomponent signals. In some embodiments, the DSP block 702 may beseparate from the VSGs 704 a-c, as shown in FIG. 7. In otherembodiments, a respective one of the parallel processing paths of theDSP block 702 may be included in each of the VSGs 704 a-c. For example,blocks 802 a, 804 a, 806 a, and 808 a may be included in VSG 704 a;blocks 802 b, 804 b, 806 b, and 808 b may be included in VSG 704 b; andblocks 802 c, 804 c, 806 c, and 808 c may be included in VSG 704 c.

In some embodiments, each of the frequency shift blocks 802 a-c mayreceive a copy of the entire digital input signal. In other embodiments,each of the frequency shift blocks 802 a-c may receive only a respectiveportion of the digital input signal comprising a respective frequencyband.

FIG. 3a illustrates an exemplary embodiment of frequency bands asreceived by the frequency shift blocks 802 a-c, represented in thefrequency domain. In this example, frequency band 302 represents thefrequency band of the frequency shift block 802 a (i.e. VSG 704 a),frequency band 304 represents the frequency band of the frequency shiftblock 802 b (i.e. VSG 704 b), and frequency band 306 represents thefrequency band of the frequency shift block 802 c (i.e. VSG 704 c). Theregion covered by frequency bands 304-306 together represents theaggregate frequency band identified by an aggregate center frequency.Each respective center frequency is offset from the aggregate centerfrequency by a respective frequency offset. In some circumstances arespective center frequency may be 0 Hz, as in the example of frequencyband 304.

The frequency bands may overlap to avoid gaps within the aggregatefrequency band. For example, the region 308 represents a region ofoverlap between frequency band 302 and frequency band 304 (i.e. betweenthe respective component signals of VSGs 704 a and 704 b), and theregion 310 represents the region of overlap between frequency band 304and frequency band 306 (i.e. between the respective component signals ofVSGs 704 b and 704 c).

At the frequency shift blocks 802 a-c, each of the respective componentsignals may be shifted to baseband. Where the aggregate frequency bandis initially at baseband, this means that each respective componentsignal is frequency-shifted by the negative of the respective frequencyoffset. For example, if the respective frequency offset of therespective component signal being processed by the frequency shift block802 c is 60 MHz, then the frequency shift block 802 c would frequencyshift the respective component signal by −60 MHz.

At the decimate blocks 804 a-c, each respective component signal may bedecimated to a rate that is less than or equal to the maximum samplerate of the corresponding VSG. This decimation may comprise merelydropping samples. Alternatively, this decimation may comprisealias-protected decimation, utilizing an alias protection filter.

The filter blocks 806 a-c are similar to the filter blocks 204 a-c ofFIG. 2. Because the respective frequency bands of the respectivecomponent signals overlap in frequency, as shown in FIG. 3, the overlapregions should be filtered to prevent power spikes in the overlapregions when the respective signals are combined by the combiner 706. Inother words, the respective component signals should be filtered suchthat their sum appears continuous. Specifically, the respectivecomponent signals may be filtered such that the sum of overlappingsignals provides a unity response at all points within the aggregatefrequency band.

Various filter shapes may be used to accomplish this. For example, FIG.4 illustrates the response of a half-band filter, where the solid trace402 represents a filter response for a first VSG and the dotted trace404 represents a filter response for a second VSG with an overlappingfrequency band. In FIG. 4, the crossover point is located at 30 MHzwhere there is a 10 MHz crossover region. While a half-band filterinherently has the needed spectral characteristics to filter overlappingfrequency bands to sum together to produce unity gain, it forces thecrossover point to occur at the sampling frequency divided by four,fs/4. In other embodiments, this crossover may be moved further out infrequency using other filter methods, e.g., to increase theeffectiveness of the natural instantaneous bandwidth of each device.

At the gain and phase correction blocks 808 a-c, the magnitude and phaseof each respective component signal may be adjusted to make the spectrumcontinuous through the regions of overlap. This gain and phasecorrection may comprise a complex multiply of each of one or more of therespective component signals with a respective calibration constant.Determining a calibration constant for each of the gain and phasecorrection blocks 808 a-c is discussed below.

Determining VSG Calibration Constants

In order to adjust the magnitudes and phases of the respective signalsin the transmit path to provide continuity through the regions ofoverlap, the relative magnitudes and phases between the respectivesignals without adjustment may be determined using one or morecalibration tones. This may be performed in multiple ways.

For example, to determine calibration constants for a system such as thesystem 700 shown in FIGS. 7-8, the system 700 may be set to acalibration mode, which may operate according to one of the followingapproaches.

In a first approach, a calibration tone may be added to the input TXSignal within a region of overlap of the respective frequency band ofthe first VSG and the respective frequency band of a second, adjacentVSG. The calibration tone may be generated by a digital calibration tonegenerator (not shown), which may be comprised within the DSP block 702,or may be a separate component. The calibration tone may be added to theinput TX Signal using a switch, a multiplexer, or by using any othermethod known in the art. Preferably, the calibration tone may begenerated at the center of the region of overlap. For example, thecalibration tone generator may generate a calibration tone 312 withinthe region 308, as shown in FIG. 3b . The respective outputs of the VSG704 a and the VSG 704 b will thus each include a representation of thecalibration tone. With the output of the VSG 704 b turned off ordisabled, the magnitude and phase of the representation of thecalibration tone 312 present in the output of the VSG 704 a may bemeasured by the calibration receiver 710. The output of the first VSGmay then be turned off or disabled, and the output of the second VSG maybe enabled. For example, the VSG 704 b may be enabled. Therepresentation of the calibration tone 312 present in the output of theVSG 704 b may then be measured by the calibration receiver 710. Wherethe calibration receiver 710 is phase-locked with the VSGs 704 a and 704b throughout the time when the two measurements are made, a relativephase difference of the VSGs 704 a and 704 b may be determined bycomparing the phases of the two representations of the calibration tone312, as measured by the calibration receiver 710. A relative magnitudedifference of the VSGs 704 a and 704 b may also be determined bycomparing the magnitudes of the two representations of the calibrationtone 312. A calibration constant may then be determined for one or moreof the VSGs 704 a and 704 b, based on the determined relative phasedifference and the determined relative magnitude difference. This methodmay be repeated for each of the regions of overlap.

In a second approach, an iterative method may be used to determine therelative magnitude and phases. In this method, both the VSG 704 a andthe VSG 704 b may each simultaneously generate an output comprising arepresentation of the calibration tone 312. Then, the magnitude andphase of the VSG 704 b may be iteratively adjusted, seeking to force therespective representations of the calibration tone 312 present in therespective outputs of VSG1 and VSG2 to deconstructively interfere. Inother words, the magnitude and phase of the representation of thecalibration tone 312 present in the output of the VSG 704 b may beiteratively adjusted until the total output power at the frequency ofthe representation of the calibration tone 312 is minimized. Thecalibration constant may then be determined by negating the VSG 704 bresult to rotate it by 180 degrees. This procedure may then be repeatedfor each overlap region. While this second approach would take longerthan the direct measurement of the first approach, this second approachdoes not require the calibration receiver 710 to be phase locked to theVSGs. Moreover, since the calibration receiver 710 is only makingunlocked power measurements, the calibration receiver 710 may bereplaced by a power meter, thereby simplifying the hardware requirementsof the calibration circuitry.

In a third approach, respective calibration tones from the VSG 704 a andthe VSG 704 b may be at different frequencies but with some knownrelationship. In this case the VSG 704 a and the VSG 704 b may eachgenerate a respective calibration tone at the same time in asynchronized manner, and the magnitude and phase differences may besimultaneously sampled. For example, FIG. 3c illustrates calibrationtone 312 a, which may be generated by the VSG 704 a, and calibrationtone 312 b, which may be generated by the VSG 704 b. Similarly,calibration tones 314 a and 314 b may be generated by the VSG 704 b andthe VSG 704 c, respectively. The complex calibration constant may becomputed by taking into consideration the known relationship between thecalibration tones. For example, a calibration constant may be determinedfor one or more of the VSGs 704 a and 704 b, based on the knownrelationship between the calibration tones 312 a and 312 b.

The calibration constants for each of the VSGs should be determined, orre-determined, each time the phases of the VSGs change relative to eachother. For example, the relative phases of the VSGs may change if therelative phases of the LOs of the VSGs change. This may occur, e.g., ifthe LO frequencies of one or more VSGs change and the one or more VSGsare relocked. In some VSGs, the phase of the LO can be made to bedeterministic. If this is the case, then a calibration constant may bedetermined once and stored for each frequency of the VSG. The storedcalibration constant may then be recalled at some future time withoutthe need for recalibration.

Although the embodiments above have been described in considerabledetail, numerous variations and modifications will become apparent tothose skilled in the art once the above disclosure is fully appreciated.It is intended that the following claims be interpreted to embrace allsuch variations and modifications.

1. A method for processing a digital signal, the method comprising:frequency-shifting each of a plurality of component signals, eachcomponent signal comprising a respective frequency band of interest ofthe digital signal, each frequency band of interest having a region ofoverlap with a frequency band of interest of at least one othercomponent signal, such that a respective center frequency of eachfrequency band of interest is shifted to baseband; shaping a frequencyresponse of the component signals by a digital filter; converting atleast the frequency bands of interest of the component signals torespective analog signals using respective signal processing pathways,wherein the component signals are phase-locked; and combining therespective analog signals to obtain a composite signal, wherein a shapeof the digital filter has a reduced gain in at least a portion of theregions of overlap to reduce artificial increases in magnitude in thecomposite signal.
 2. The method of claim 1, wherein converting afrequency band of interest to a respective analog signal comprisessampling at least a portion of the digital signal corresponding to thefrequency band of interest.
 3. The method of claim 1, further comprisingadjusting at least one of gain and phase of the component signals, suchthat the composite signal has a continuous frequency response over anaggregate frequency band.
 4. The method of claim 3, wherein saidconverting at least the frequency bands of interest of the componentsignals is configured to cause an aggregate center frequency of theaggregate frequency band to be located at a desired carrier frequency,and each respective center frequency of a respective frequency band ofinterest to be offset from the desired carrier frequency by a respectivefrequency offset by which the respective frequency band of interest wasshifted during said frequency-shifting.
 5. The method of claim 1,further comprising decimating at least the frequency bands of interestof the component signals to a rate that is less than or equal to amaximum sample rate of the signal processing pathways.
 6. The method ofclaim 5, further comprising adjusting at least one of gain and phase ofthe component signals, such that the composite signal has a continuousfrequency response over the aggregate frequency band, wherein saidshaping a frequency response of the component signals is performed aftersaid decimating and before said adjusting at least one of gain andphase.
 7. The method of claim 6, wherein the digital filter is ahalf-band filter.
 8. An apparatus for processing a digital signal, theapparatus comprising: a plurality of parallel signal processingpathways, each of the parallel signal processing pathways configured to:frequency-shift a respective component signal comprising a respectivefrequency band of interest of the digital signal, the frequency band ofinterest having a region of overlap with a frequency band of interest ofat least one other component signal, such that a center frequency of thefrequency band of interest is shifted to baseband; shape a frequencyresponse of at least the respective frequency band of interest, using adigital filter; and convert at least the frequency band of interest to arespective analog signal, wherein the component signals arephase-locked; and a signal combiner configured to combine the respectiveanalog signals to obtain a composite signal, wherein a shape of thedigital filter has a reduced gain in at least a portion of the regionsof overlap to reduce artificial increases in magnitude in the compositesignal.
 9. The apparatus of claim 8, wherein each of the parallel signalprocessing paths is further configured to: decimate at least therespective frequency band of interest to a rate that is less than orequal to a maximum sample rate of the signal processing pathway.
 10. Theapparatus of claim 8, wherein each of the parallel signal processingpathways is further configured to: adjust gain and phase of at least therespective frequency band of interest, wherein adjusting the gain andphase in each of the parallel signal processing pathways is configuredto cause the composite signal to have a continuous frequency responseover an aggregate frequency band.
 11. The apparatus of claim 10, whereineach of the parallel signal processing paths is further configured to:decimate at least the respective frequency band of interest to a ratethat is less than or equal to a maximum sample rate of the signalprocessing pathway, wherein said shaping a frequency response of atleast the respective frequency band of interest is performed after saiddecimating and before said adjusting gain and phase.
 12. The apparatusof claim 11, wherein the digital filter is a half-band filter.
 13. Theapparatus of claim 8, wherein said converting at least the frequencyband of interest in each of the parallel signal processing pathways isconfigured to cause an aggregate center frequency of an aggregatefrequency band to be located at a desired carrier frequency, and eachrespective center frequency of a respective frequency band of interestto be offset from the desired carrier frequency by a respectivefrequency offset by which the respective frequency band of interest wasshifted during said frequency-shifting.
 14. An apparatus for processinga digital signal, the apparatus comprising: a first signal processingpathway configured to: receive a first component signal comprising afirst frequency band of the digital signal; and convert at least thefirst frequency band to a first analog signal; a second signalprocessing pathway configured to: receive a second component signalphase-locked with the first component signal, the second componentsignal comprising a second frequency band of the digital signal, thesecond frequency band having a region of overlap with the firstfrequency band; and convert at least the second frequency band to asecond analog signal; a signal combiner configured to combine the firstanalog signal and the second analog signal to obtain a composite signal;a calibration receiver configured to, while the apparatus is in acalibration mode, compute a calibration constant useable to correctphase mismatch between the first signal processing pathway and thesecond signal processing pathway, wherein the calibration constant iscomputed based on a phase difference between a first version of acalibration tone output by the signal combiner while the first signalprocessing pathway is disabled, and a second version of the calibrationtone output by the signal combiner while the second signal processingpathway is disabled, wherein the calibration tone is included in thedigital signal within the region of overlap between the first frequencyband and the second frequency band; and a memory configured to store thecalibration constant.
 15. The apparatus of claim 14, wherein, when theapparatus is not in the calibration mode, the second signal processingpathway is further configured to: correct phase mismatch between anoutput of the second signal processing pathway and an output of thefirst signal processing pathway, using the stored calibration constant.16. The apparatus of claim 14, wherein the calibration constant isfurther useable to correct gain mismatch between the first signalprocessing pathway and the second signal processing pathway, wherein thecalibration constant is computed further based on a ratio of a magnitudeof the first version of the calibration tone and a magnitude of thesecond version of the calibration tone.
 17. The apparatus of claim 14,wherein the first signal processing pathway is further configured toshape a frequency response of the first component signal by a firstdigital filter; wherein the second signal processing pathway is furtherconfigured to shape a frequency response of the second component signalby a second digital filter; and wherein the first digital filter and thesecond digital filter have reduced gain in at least a portion of theregion of overlap to reduce artificial increases in the compositesignal.
 18. The apparatus of claim 17, wherein the first digital filterand the second digital filter are half-band filters.
 19. The apparatusof claim 14, wherein the first signal processing pathway is configuredto shift the first frequency band to baseband; wherein the second signalprocessing pathway is configured to shift the second frequency band tobaseband; wherein said converting at least the first frequency bandcomprises frequency-shifting the first analog signal such that thecalibration tone occurs at a particular frequency, and said convertingat least the second frequency band comprises frequency-shifting thesecond analog signal such that the calibration tone occurs at theparticular frequency.
 20. The apparatus of claim 19, wherein the firstsignal processing pathway is further configured to decimate the firstcomponent signal to a rate that is less than or equal to a maximumsample rate of the first signal processing pathway, wherein saiddecimating the first component signal is performed after said shiftingthe first frequency band to baseband and before said converting thefirst frequency band; and wherein the second signal processing pathwayis further configured to decimate the second component signal to a ratethat is less than or equal to a maximum sample rate of the second signalprocessing pathway, wherein said decimating the second component signalis performed after said shifting the second frequency band to basebandand before said converting the second frequency band.